EDN Network

Nexperia has expanded its GaN FET portfolio with 12 new E-mode devices, available in both low- and high-voltage options. The additions address the demand for higher efficiency and compact designs across consumer, industrial, server/computing, and telecommunications markets. Nexperia’s portfolio includes both cascode and E-mode GaN FETs, available in a wide variety of packages, providing flexibility for diverse design needs.

The new offerings include 40-V bidirectional devices (RDS(on) <12 mΩ), designed for overvoltage protection, load switching, and low-voltage applications such as battery management systems in mobile devices and laptop computers. These devices provide critical support for applications requiring efficient and reliable switching.

Also featured are 100-V and 150-V devices (RDS(on) <7 mΩ), useful for synchronous rectification in power supplies for consumer devices, DC/DC converters in datacom and telecom equipment, photovoltaic micro-inverters, Class-D audio amplifiers, and motor control systems in e-bikes, forklifts, and light electric vehicles. The release also includes 700-V devices (RDS(on) >140 mΩ) for LED drivers and power factor correction (PFC) applications, along with 650-V devices (RDS(on) >350 mΩ) suitable for AC/DC converters, where slightly higher on-resistance is acceptable for the specific application.

To learn more about Nexperia’s E-mode GaN FETs, click here.

Nexperia

Find more datasheets on products like this one at Datasheets.com, searchable by category, part #, description, manufacturer, and more.

The post Broad GaN FET lineup eases design headaches appeared first on EDN.

NVIDIA’s Spectrum-X and Quantum-X silicon photonics-based network switches connect millions of GPUs, scaling AI compute. They achieve up to 1.6 Tbps per port and up to 400 Tbps aggregate bandwidth. NVIDIA reports the switch platforms use 4x fewer lasers for 3.5x better power efficiency, 63x greater signal integrity, 10x higher network resiliency at scale, and 1.3x faster deployment than conventional networks.

Spectrum-X Photonics Ethernet switches support 128 ports of 800 Gbps or 512 ports of 200 Gbps, delivering 100 Tbps of total bandwidth. A high-capacity variant offers 512 ports of 800 Gbps or 2048 ports of 200 Gbps, for a total throughput of 400 Tbps.

Quantum-X Photonics InfiniBand switches provide 144 ports of 800 Gbps, achieved using 200 Gbps SerDes per port. Built-in liquid cooling keeps the onboard silicon photonics from overheating. According to NVIDIA, Quantum-X Photonics switches are 2x faster and offer 5x higher scalability for AI compute fabrics compared to the previous generation.

NVIDIA’s silicon photonics ecosystem includes collaborations with TSMC, Coherent, Corning, Foxconn, Lumentum, and SENKO to develop an integrated silicon-optics process and robust supply chain.

Quantum-X Photonics InfiniBand switches are expected to be available later this year. Spectrum-X Photonics Ethernet switches will be coming in 2026 from leading infrastructure and system vendors. Learn more about NVIDIA’s silicon photonics here.

NVIDIA

Find more datasheets on products like this one at Datasheets.com, searchable by category, part #, description, manufacturer, and more.

The post NVIDIA switches scale AI with silicon photonics appeared first on EDN.

Frequent design idea (DI) contributor Nick Cornford recently published a synergistic pair of DIs “A pitch-linear VCO, part 1: Getting it going” and “A pitch-linear VCO, part 2: taking it further.”

Wow the engineering world with your unique design: Design Ideas Submission Guide

The main theme of these articles is design techniques for audio VCOs that have an exponential (a.k.a. linear in pitch) relationship between control voltage and frequency. Great work Nick! I became particularly interested in the topic during a lively discussion (typical of editor Aalyia’s DI kitchen) in the comments section. The debate was about whether such a VCO could be built around the venerable 555 analog timer. Some said nay, others yea. I leaned toward the latter opinion and decided to try to put a schematic where my mouth was. Figure 1 is the result.

Figure 1 555 VCO discharges timing cap C1 completely to the negative rail via a Reset pulse.

The nay-sayers’ case hinged on a perceived inability of the 555 architecture to completely discharge the timing capacitor, C1 in Figure 1. They seemed to have a good argument because, in its usual mode of operation, the discharge of C1 ends when the trigger input level is crossed. This normally happens at one third of the supply rail differential and one third is a long way from zero! But it turns out the 555, despite being such an old dog, knows a different trick, it involves a very seldom used feature of this ancient chip: the reset pin 4.

The 555 datasheet says a pulse on reset will override trigger and also force discharge of C1. In Figure 1, R3 and C2 provide such a pulse when the OUT pin goes low at the end of the timing cycle. The R3C2 product ensures the pulse is long enough for the 15 Ω Ron of the Dch pin to accurately evacuate C1. 

And that’s it. Problem solved as sketched in Figure 2.

Figure 2 The VCO waveforms; reset pulses at the end of each timing cycle, and is triggered when Vc1 = Vcon, to force an adequately complete discharge of C1.

Figure 3 illustrates the resulting satisfactory log conformity (due mostly to my shameless theft of Nick’s clever resistor ratios) of the resulting 555. VCO, showing good exponential (linear in pitch) behavior over the desired two octaves of 250 to 1000 Hz.

Figure 3 Log plot of the frequency versus control voltage for the two-octave linear-in-pitch VCO. [X axis = Vcon volts (inverted), Y axis = Hz / 16 = 250 Hz to 1 kHz]

In fact, at the price of an extra resistor, it might be possible to improve linearity enough to pick up another half a volt and half an octave on both ends of the pitch range to span 177 Hz to 1410 Hz. See Figure 4 and Figure 5.

Figure 4 R4 sums ~6% of Vcon with the C1 timing ramp to get the improvement in linearity shown in Figure 5.

Figure 5 The effect of the R4 modification showing a linearity improvement. [X axis = Vcon volts (inverted), Y axis = Hz / 16]

Stephen Woodward’s relationship with EDN’s DI column goes back quite a long way. Over 100 submissions have been accepted since his first contribution back in 1974.

Related Content

• A pitch-linear VCO, part 1: Getting it going
• A pitch-linear VCO, part 2: taking it further
• VCO using the TL431 reference
• Ultra-low distortion oscillator, part 1: how not to do it.
• How to control your impulses—part 1
• A two transistor sine wave oscillator

The post Can a free running LMC555 VCO discharge its timing cap to zero? appeared first on EDN.

This year during APEC, much of the focus on the show floor revolved around data center tech, with companies showcasing high-density power supply units (PSU), battery backup units (BBU), intermediate bus converters (IBC), and GPU solutions (Figure 1). 

Figure 1: Up to 12 kW Infineon PSU technology leverages a mixture of the CoolSIC, CoolMOS, and CoolGaN technologies. 

The motivation comes from the massive power demand increase that the generative AI, in particular, LLMs have brought on, shooting up the 2% of global power consumption from data centers to a projected 7% by 2030. This power demand originates from the shift from the more 120 kV (single-phase AC) stepped down to 48 V to 250-350 kV (three-phase AC) stepped down to 400 VDC rails attached to the rack and distributed from there (to switches, PSUs, compute trays, switch trays, BBUs, and GPUs).

Infineon’s booth presented a comprehensive suite of solutions from the “power grid to the core.” The BBU technology (Figure 2) utilizes the partial power converter (PPC) topology to enable high power densities (> 12 kW) using scalable 4 kW power converter cards.

Figure 2: Infineon BBU roadmap, using both Si and GaN to scale up the power density of the converters with high efficiencies. Source: Infineon

The technology boasts an efficiency of 99.5% using lower voltage (40 V and 80 V) switches to increase figure of merit (FOM) and yield efficiency gains. The solutions are aimed at meeting space-restrictions of modern BBUs that are outfitted with more and more batteries and hence less space for the embedded DC/DC converter.

Their latest generation of vertical power delivery modules feature a leap in GPU/AI card power delivery, offering up to 2 A/mm2. These improvements create massive space-savings on the already space-constrained AI cards that often require 2000 A to 3000 A for power-hungry chips such as the Nvidia Blackwell GPU.

Instead of being mounted laterally, or alongside the chip, these devices deliver power on the underside of the card to massively reduce power delivery losses. The backside mounting does come with its profile restraints; there is a max height of 5 mm to facilitate heatsink mounting on the other side of the board, so these modules must maintain their 4-mm height. 

The first generation of the dual-phase module featured the silicon device that sat on top of the substrate with integrated inductors and capacitors to achieve 1 A/mm2, or 140A max,  in a 10 x 9 mm package. This was followed by a dual-phase module that featured a 1.5 A/mm2, or 160 A max, improvement within 8 x 8 mm dimensions. Embedding the silicon into the substrate to have only one PCB is what contributed to the major space-savings in this iteration (Figure 4). 

Figure 4: The second generation of Infineon vertical power delivery modules mounted on the backside of GPU PCB deliver a total of 2000 A. An Infineon controller IC can also be seen providing the necessary voltage/current through coordination with the vertical power delivery modules and chip.

The third generation just released has brought on two more power stages for a quad-phase module for 2 A/mm2, or 280 A max, in the 10 x 9 mm space; doubling the current density of the first generation in the same space (Figure 5). 

Figure 5: Third generation of Infineon vertical power delivery modules are mounted on the backside of GPU PCB delivering a total of 2,000 A. 

Custom solutions can go beyond this, integrating more power stages in a single substrate. Other enhancements include bypassing the motherboard and direct-attaching to the substrate in the GPU since PCB substrate materials are lossy for signals with high current densities.

However, this calls for closer collaboration with SoC vendors that are willing to implement system-level solutions. High current density solutions are in the works with Infineon, potentially doubling the current density with another multi-phase module.

The Navitas booth also showed two form factors of PSUs: a common redundant power supply (CRPS) form factor and a longer PSU that meets open compute project (OCP) guidelines and compiled to the ORv3 base specification (Figure 6). The CRPS solution delivers 4.5 kW with two-stages including a SiC PFC end and GaN LLC and offers titanium level efficiency.

Figure 6: Typical rack is shown with RAM, GPU, PSUs, and airflow outlet with barrel fans. The PSUs conform to the CRPS and provide redundancy to encourage zero downtime in the event of transient faults, brownouts, and blackouts.

Hyperscalers or high performance compute (HPC) applications that utilize the OCP architecture can install PSUs in a row to centralize power in the rack. The Navitas PSU offered for this datacenter topology offers up to 8.5 kW with up to a 98% efficiency using a three-phase interleaved CCM totem pole SiC PFC and three-phase GaN LLC (Figure 7).

Figure 7: Navitas 8.5 kW PSU is geared toward hyperscalers using both Gen-3 Fast SiC and GaNSafe devices.

Aalyia Shaukat, associate editor at EDN, has worked in the design publishing industry for six years. She holds a Bachelor’s degree in electrical engineering from Rochester Institute of Technology, and has published works in major EE journals as well as trade publications.

Related Content

• Data center power meets rising energy demands amid AI boom
• Telecom/Datacom 48V: Have no fear; 400V is here
• Power Tips #139: How to simplify AC/DC flyback design with a self-biased converter
• Data center next generation power supply solutions for improved efficiency

The post Data center solutions take center stage at APEC 2025 appeared first on EDN.

My recent teardown of a rechargeable vape device was…wow…popular. I suspected upfront that it might cultivate a modicum of incremental traffic from the vape-using (and vape-curious) general public, but…wow. This long-planned follow-up focuses on non-rechargeable (i.e., disposable) vape counterparts, in part fueled by my own curiosity as to their contents but more generally and predominantly driven by my long-standing bleeds-green environmental outrage.

Here’s an example of what I mean, showcased in a recent Slashdot post that highlighted a writeup in The Guardian:

Thirteen vapes are thrown away every second in the UK — more than a million a day — leading to an “environmental nightmare,” according to research.

There has also been a rise in “big puff” vapes which are bigger and can hold up to 6,000 puffs per vape, with single use vapes averaging 600. Three million of these larger vapes are being bought every week according to the research, commissioned by Material Focus, and conducted by Opinium. 8.2 million vapes are now thrown away or recycled incorrectly every week.

From June 2025 it will be illegal to sell single-use vapes, a move designed to combat environmental damage and their widespread use by children. Vapes will only be allowed to be sold if they are rechargeable or contain a refillable cartridge.

But all types of vape contain lithium-ion batteries which are dangerous if crushed or damaged because they can cause fires in bin lorries or waste and recycling centres. These fires are on the rise across the UK, with an increase last year of 71% compared with 2022.

I have (at least) two questions:

• If there are a million toxic chemical- and metal-leaching vapes headed to landfills (if we’re lucky; many, more likely, are sent directly into the water table via casual, irresponsible discard wherever it’s convenient for the owner to toss ‘em) in the UK alone, what’s that number look like when extrapolated to a worldwide count? Truthfully, from a blood pressure standpoint, I’m not sure I want to know the answer to that one.
• And why are vapes that are “rechargeable or contain a refillable cartridge” (bolded emphasis mine) excluded from the upcoming UK ban? Why can’t (and shouldn’t) it instead be only those that are “rechargeable and contain a refillable cartridge”?

Rant off. One of the comments I posted as follow-up to last November’s initial entry in this vape-teardown series pointed readers to near-coincident published related coverage in Ars Technica:

Disposable vapes are indefensible. Many, or maybe most, of them contain rechargeable lithium-ion batteries, but manufacturers prefer to sell new ones. More than 260 million vape batteries are estimated to enter the trash stream every year in the UK alone. Vapers and vape makers are simply leaving an e-waste epidemic to the planet’s future residents to sort out.

To make a point about how wasteful this practice is—and to also make a pretty rad project and video—Chris Doel took 130 disposable vape batteries (the bigger “3,500 puff” types with model 20400 cells) found littered at a music festival and converted them into a 48-volt, 1,500-watt e-bike battery, one that powered an e-bike with almost no pedaling more than 20 miles.

The accompanying video is well worth your viewing time, IMHO.

and gave me the confidence to attempt my own teardown of conceptually similar vape devices, since Doel had confidently just ripped off the tip and back ends to get to their insides. Here’s the implement of destruction that I personally used:

And here are today’s victims, extracted from the trash as was the case with their rechargeable predecessor, and as usual accompanied by a 0.75″ (19.1 mm) diameter U.S. penny for size comparison purposes (not to mention roll-away prevention purposes):

The upper one is actually (supposedly, although there are still loopholes, apparently) no longer available in the US. It’s the “4000 puff” Noms X product variant (and Mojito Mint flavor) of the Esco Bars brand, manufactured by the Chinese company Shenzhen Innokin Technology. And no, I have no idea what “Pastel Cartel” means. The lower vape is Mr Fog ‘s “2000 puffs” Max pro model (and Raspberry Grape Black Currant flavor).

Here are their respective tips:

And their bottoms:

The black-color bottom end of the Esco Bars vape is fixed in position; note the two holes for incoming-airflow purposes. You’ll shortly see what secondary function the one in the middle also serves; that said, I’m not sure of the purpose of the incremental smaller second offset one. The white-color end of the Mr Fog vape, conversely, can be rotated to user-adjust the airflow. The two vents are on the sides of the end piece; here’s how airflow adjustment operates:

and briefly jumping ahead in time mid-teardown, here’s how it’s implemented:

Let’s start the disassembly process with the Esco Bars device, as previously mentioned by wrenching the bottom piece off with my pliers (see what I did there?).

That black rectangular spongy piece went flying when I pulled the bottom piece off, but I’m guessing from the lingering indentations that it normally sits in-between that thing that looks like a microphone (and fits inside the circular middle portion of the bottom piece) and the battery. And about that “thing that looks like a microphone”…I was initially a bit flummoxed when I saw it (no, I never thought it was actually a microphone, although other folks were amusingly-to-me apparently convinced otherwise), until I realized that neither vape has an on-off switch. Instead, what you do to “turn them on” (i.e., power up the heating coil) is to suck on the tip, which vapers refer to as a “draw”.

This “thing that looks like a microphone”, apparently, is a “draw sensor”; it detects the resultant user-generated airflow that’s initiated from the bottom and (as is already obvious even with the battery still in place) passes from there through the gap between the battery and vape body. This Quora thread has all the details, including pictures of a sensor that looks just like the one in the Esco Bars vape (and the Mr Fog one, for that matter, prematurely ruining the surprise…sorry). I’m guessing that the red and black wires route to the sensor from the battery, and the blue one carries a signal sent by the sensor to the heating coil when airflow is detected.

By repeatedly shaking the vape device (with a foam cushion underneath, in case the contents went flying) I got the battery out of the case far enough:

that I was then able to get a grip on it with my fingers and pull it the rest of the way out:

The remainder of the internals remained stubbornly stuck at the rear end of the tube until I started twisting on the tip with the wrench:

At which point the translucent tube fell out the bottom, too. Disgusting (and oily, too), huh?

From my research (I’ve learned more than I ever wanted to about vapes the past 24 hours or so), inside the plastic tube are apparently nicotine salts, soaked in the flavored vape juice. Here’s the entirety of the insides, stretched out:

And here’s what you’ve all been waiting for, the battery specs, 3.7V and 5.55 Wh:

Now for the Mr Fog vape. Again, I started with the white bottom piece, which initially didn’t get me very far (although look; another “microphone”):

So, I switched to the tip, which didn’t get me much further along…and yuck, again:

Back to the bottom for more twisting, this time of the clear plastic piece that as I showed you earlier, the white bottom piece fits around. That’s better:

Once again, a combination of shaking and two-finger pinching-and-pulling got the battery out:

But this time I had to then push from the top to get the rest out:

Greasy, smelly mission completed:

And the battery specs: once again 3.7V, but this time only 4.07 wH/1100 mAh, reflective of the Mr Fog vape’s comparative “half the puffs” estimate versus the Esco Bars alternative.

In closing, what most surprised me, I guess, is that neither of these vapes use standard 18650 cells found in a diversity of other devices (although from some of my research, their limited spec’d peak output current capabilities might be a coil-heating hinderance or, worse, a thermal safety complication in this particular application), or even the less common 20400 ones showcased in the video at the beginning of this writeup. With that, I’ll wrap up, take a deep draw (of nicotine-free air, mind you) and await your thoughts in the comments!

—Brian Dipert is the Editor-in-Chief of the Edge AI and Vision Alliance, and a Senior Analyst at BDTI and Editor-in-Chief of InsideDSP, the company’s online newsletter.

Related Content

• Investigating a vape device
• SLA batteries: More system form factors and lithium-based successors
• Introduction to lithium-ion rechargeable battery design
• Diagnosing and resuscitating a set of DJI drone batteries

The post Disposable vapes: Unnecessary, excessive waste in cylindrical shapes appeared first on EDN.

Texas Instruments’ APEC-related releases are power management chips centered around supporting the AI-driven power demands in data centers. The releases include the first 48-V integrated hot-swap eFuse with power-path protection (TPS1685) and an integrated GaN power stage (gate driver + FET) in the industry-standard TOLL package. 

In a conversation with Priya Thanigai, VP and Business Unit Manager of power switches at Texas Instruments, EDN obtained some insights on meeting the needs of next-generation racks demanding the 48-V architecture.

Spotlight on data centers

Hot topics at APEC typically encompassed the use of wide bandgap semiconductors like silicon carbide (SiC) and gallium nitride (GaN) to yield higher efficiency subsystems in the steady electrification of technologies. Electrified end applications have spanned from e-mobility to industrial processes that are enabled by battery and smart grid advancements. 

Discussions this year have shifted more toward the power demands that generative AI has created for data centers. While much of the actual power consumption of these data centers remains secretive, it’s apparent that LLMs like ChatGPT and DeepSeek have created a substantial increase; the U.S. data center electricity usage tripled from 2014 to 2023 according to the U.S. department of energy (DoE). The number is anticipated to double or triple by 2028.

The international energy agency (IEA) also reported that data centers consumed ~1.4-1.7% of global electricity in 2022; this is also expected to double by 2026. According to the World Economic Forum, “the computational power needed for sustaining AI’s growth is doubling roughly every 100 days.”

Going nuclear

Hyperscalers are also making more apparent their plans to sustain the energy demands. In September 2024, plans to recommission the Three Mile Island nuclear plant were made public with a 20-year contract to help power Microsoft data centers. Other technology companies follow a similar nuclear path, augmenting power capabilities with small modular reactors (SMRs).

And as the semiconductor industry is feverishly fabricating chips that can efficiently run these compute-intensive training tasks through software-hardware codesign, the power demands continually soar. Further into the future, these nuclear reactors could be used with solid-state transformers to support data center processing.

The 48-V bus and beyond

The data center server room consists of a sea of IT racks supported by a sidecar that holds hot-swappable power supply units (PSUs) that facilitate replacing or upgrading a PSU without shutting down the server (Figure 1). These PSUs support much higher power densities moving from 6 kW with the 48-V bus to 100 MW with the 400-V bus.

Figure 1: Sidecar, IT rack, and supporting subsystems shown at the TI booth during APEC 2025. 

“While data centers have been ahead of the curve, cars are only now moving to 48 V,” said Thanigai. “But data centers have probably already been there for about a decade.” It’s just been very slow because earlier systems really didn’t need the compute power until LLMs exploded. Until then, it was only the high-end GPUs that needed that extra power at 48 V.

She mentioned how TI had been keeping a watchful eye on the relatively slow move from 12-V products for data centers 48-V and how recent pressures have brought on that inflection point. “Now we’re seeing more native 48-V systems ship and we’re talking about 400-V already,” Thanigai said. “So the transition from 12 V to 48 V may have taken a decade to hit the inflection point but 48 V to 400 V will probably be shorter and sharper because of how much energy is needed by data centers.”

Moving from discretes to integrated eFuses

Power path protection is tied directly to PSU reliability and is therefore a critical aspect of ensuring zero downtime deployments. The 48-V eFuse is a successor to the popular 12-V eFuse category; the shift to 48 V allows users to scale power to beyond 6 kW. 

“If you’re looking at the power design transition, generally power architectures will begin with discretes at the start of any design because they want to get a good feel of how to build something,” explained Thanigai. The building blocks of power path protection generally include the power FET, a gate or voltage drive to drive it, and components like a soft-start capacitor to control the inrush, comparators, and current-sense elements.

Thanigai described the moves toward more integration where the hot swap controller integrates the amplifiers, some of the protection features, and some of the smarts. However, there still remains an external FET and sensing element. 

“The last leg of the integration is eFuse where the FET, the controller, and all the smarts are in a single chip,” she said. “That’s a classic power design evolution, where you go from discrete to semi-integrated to fully integrated.” The TPS1685 eFuse includes protection features like rapid response to fault events with an integrated black box for fault logging. Then there is a user-configurable overcurrent blanking timer that avoids false tripping at peak inrush.

Advanced stacking for loads > 6 kW

Mismatches in the on-state resistance (Rdson) due to PCB trace resistance and comparator thresholds can create false tripping (Figure 2). The conventional discrete designs require power architects to hand calculate the margins to make sure the FETs are matched such that no single FET is taking on more thermal stress than the others.  

Figure 2: Discrete implementations require individual calculations per sense element and FET to take into account mismatches at each node; instead Rdson is actively adjusted via Vgs regulation and equal steady-state current across all devices is achieved through path resistance equalization. Source: Texas Instruments 

The IP in the TPS1685 eFuse actively measures and monitors the thermal stress at various areas of the FET within each of the eFuses and balances current between each automatically through a single-wire protocol. The integration designates one eFuse as the primary controller to monitor total system current by using the interconnected IMON pins, enabling active RDS(ON) shifting to ensure devices are current-sharing.

“You can basically stack unlimited eFuses,” said Thanigai, “We’ve shown up to 12 operational eFuses on a customer board and each of them can do 1 kW (~ 50 V @ 20 A), so we easily reach the 5-10 kW that you see with systems nowadays. But we can scale higher than that since there’s no upper limit.”

Figure 3: Image of 6 eFuses stacked in parallel on the top and bottom of a PCB to support a maximum load current of 120 A. 

Moving toward 400 V

When asked about the move toward supporting 400-V bus architectures, Thanigai responded, “There’s two aspects in these eFuses.” There’s the pure analog power domain, which is the FET architectures, and then there’s the digital domain which embodies smarts around the FET, she added.

All of the digital IP TI has developed scales from 12 V to 48 V to 400 V, and that while this particular device includes 48-V power FETs, TI is preparing to scale this up to 400 V.

Aalyia Shaukat, associate editor at EDN, has worked in the design publishing industry for six years. She holds a Bachelor’s degree in electrical engineering from Rochester Institute of Technology, and has published works in major EE journals as well as trade publications.

Related Content

• Telecom/Datacom 48V: Have no fear; 400V is here
• Power Tips #139: How to simplify AC/DC flyback design with a self-biased converter
• CES 2025: Moving toward software-defined vehicles
• Data center next generation power supply solutions for improved efficiency

The post Data center power meets rising energy demands amid AI boom appeared first on EDN.

The venerable Stephen Woodward recently published the design idea (DI) “Flip ON flop OFF” that converts a momentary push button to a classic push-on, push-off switch. Figure 1 is an attempt to go further still in terms of economy.

The circuit shown in Figure 1 utilizes only one half of a dual D-type package and one more capacitor to the original parts count. It also incorporates an RC power on set (or reset), to guarantee the initial state of the switch when power is applied.

Figure 1 U1A debounces SW1 via R1 & C2 so U1A can reliably toggle.

Wow the engineering world with your unique design: Design Ideas Submission Guide

The initial state of the switch is determined by the Set pin of U1A following the rising voltage on the power input due to the initial discharged state of C1. Capacitor C1 then charges towards ground leaving the flip-flop with the Q output high and the PMOS off.

Alternatively, this RC power on Set circuit can be wired to the Reset pin to change the initial power on state of the switch. The device ESD clamping diodes provide the capacitor discharge path when power is turned off.

The D-type flip-flop is essentially connected in the familiar way of Q-bar to D-input to form a bistable with each clock rising edge toggling the output state. However, in this case the combination of R1 and C2 form a delay network which prevents rapid changes on the D-input, thus effectively de-bouncing the switch by inhibiting state changes until C2 has charged/discharged to the state on the Q-bar output.

—Chris Nother built a discrete Tx/Rx for model aircraft at an early age, later discovering the dreaded “Mains Hum” in a home built “Dinsdale” Hi-Fi amplifier. Employed in R&D using the then newly available available CMOS logic from Motorola and Nat-Semi, career changes to Mainframe Computers, design of disk drive automated test equipment and storage solutions, finally turning full circle in retirement to the hobby that started it all.

Related Content

• Flip ON flop OFF
• Use an op amp as a set/reset flip-flop
• New VFC uses flip-flops as high speed, precision analog switches
• CMOS flip-flop used “off label” implements precision capacitance sensor

The post Latching D-type CMOS power switch: A “Flip ON Flop OFF” alternative appeared first on EDN.

Introduction

The demand for smaller, lighter, and more efficient AC/DC USB power delivery (PD) chargers is always a challenge for power-supply design engineers. Below 100 W, the quasi-resonant flyback is still the dominating topology, and gallium nitride (GaN) technology can push the power density and efficiency further.

However, providing bias power for the primary controller requires an auxiliary winding on the transformer as well as rectifying and filtering circuitry. To make things worse, the USB PD charger output voltage has a wide range. For example, the USB PD standard power range covers output voltages from 5 V to 20 V, and the latest USB PD extended power range allows the output voltage to go as high as 48 V. Since the auxiliary voltage is proportional to the output voltage, the bias voltage range on the primary controller will increase, requiring extra circuitry and degrading efficiency. In this power tip, I’ll introduce a self-biased flyback converter solution to address these design challenges.

Dealing with wide bias voltages

Figure 1, Figure 2, Figure 3, and Figure 4 show four different ways to deal with the wide bias voltage range in USB PD charger applications. Conventional methods include using a linear regulator, a tapped auxiliary winding, or even adding an extra DC/DC switching converter to regulate the bias voltage. All of these methods will increase component count, add cost, or increase power losses. Alternatively, self-biasing totally removes external components and increases efficiency.

Figure 1 Bias circuits for applications with wide output voltage ranges using a discrete linear regulator. Source: Texas Instruments

Figure 2 Bias circuits for applications with wide output voltage ranges using a tapped auxiliary winding. Source: Texas Instruments

Figure 3 Bias circuits for applications with wide output voltage ranges using boost converter. Source: Texas Instruments

Figure 4 Bias circuits for applications with wide output voltage ranges using a self-biased VCC. Source: Texas Instruments

VCC self-biasing

The flyback controller can always get bias power directly from the rectified AC input voltage, but this results in excessive power losses. The key to self-biasing is to harvest energy from the power stage, which can come from two sources. One is the switch-node capacitor stored energy; the other is energy stored in the primary-side winding of the transformer. As shown in Figure 5, an integrated self-biasing circuit can ideally do both, based on the input and output conditions.

Figure 5 The self-bias circuit harvests energy from the switch-node capacitance or magnetizing inductance. Source: Texas Instruments

Figure 6 shows the energy harvesting from the switch-node capacitor. This can save efficiency as it recycles the energy storage in switching node capacitor in every switching cycle. In cases such as AC low-line input when the reflected output voltage is identical to the input voltage, natural zero voltage switching will occur, and there is no energy in the switch-node capacitor, inductor energy harvesting will take effect, where a small portion of the primary switching current is directed to the VCC cap through an internal path.

Figure 6 VCC self-bias operation: (a) capacitor energy harvesting on the switching node and (b) inductor energy harvesting through the primary current. Source: Texas Instruments

Achieving auxless sensing

Many flyback controllers use the auxiliary winding to sense the input and output voltages and detect conditions such as output overvoltage or input undervoltage. With self-biased flyback converters, it is possible to use the switching-node voltage for input and output voltage sensing. As shown in Figure 7, the sensed voltage is the sum of the input and reflected output voltage. Since the average voltage across the primary winding is zero, the average of the switch-node voltage is equal to the input voltage.

For output voltage sensing, it can sample the reflected output voltage, and the controller needs to be informed of the exact turns ratio of the transformer with the use of a resistor-programmable pin [the TR pin in the Texas Instruments (TI) UCG28826].

Figure 7 Auxless voltage sensing where the sensed voltage is the sum of the input and reflected output voltage. Source: Texas Instruments

Once properly configured, self-biased devices such as the UCG28826 can accurately provide various protections like overpower and overvoltage protection. Figure 8 shows the UCG28826 in a USB PD application.

Figure 8 A self-biased USB PD design using the UCG28826 that can accurately provide various protections like overpower and overvoltage protection. Source: Texas Instruments

Figure 9 shows the overvoltage protection waveforms after intentionally disconnecting the feedback pin which is a single fault condition. The controller senses the output voltage and triggers overvoltage protection accordingly when the output ramps up to around 24.4 V for a nominal 20 V output.

Figure 9 Auxless sensing example for overvoltage protection. Channel 1 (CH1) is Vout and channel 2 (CH2) is Vsw. Source: Texas Instruments

Prototype and test result

Figure 10 shows the TI universal AC-input 65W dual USB type-C port USB PD charger reference design with an integrated GaN power switch. Due to the simplified self-bias feature and integrated GaN switch in the UCG28826, the reference design achieves a power density of 2.3 W/cm3 and 93.2% efficiency for the AC/DC stage. The auxless design also simplifies transformer manufacturing and reduces costs. Table 1 summarizes the design parameters of 65 W design for reference. 

Figure 10 A universal AC-input 65-W reference design board. Source: Texas Instruments

Parameter	Value
AC input voltage	90-264 VAC
Output voltage and current	5-20 V, 3.25 A maximum
Transformer	ATQ23-14
Turns ratio	7-to-1
Transformer inductance	200 µH
Switching frequency (full load)	90-140 kHz
Efficiency	93.2% at 90 VAC (AC/DC stage only)
Power density	2.3 W/cm3

Table 1 Universal AC-input 65W reference design parameters.

Simplified USB PD charger

A high-level integration with a controller and GaN switch can simplify USB PD charger design, but the bias circuitry for the controller and associated auxiliary winding on the transformer are still there, degrading efficiency and affecting size and cost. An integrated self-biasing circuit can eliminate that portion of the circuit and increase the power density for power supplies with wide-range outputs. Additionally, it is still possible to achieve proper input and output voltage sensing in the absence of an auxiliary winding on the transformer.

Max Wang is a systems engineer and Member, Group Technical Staff at Texas Instruments. He has over 18 years of experience in the power semiconductor and power-supply industries in computing, industrial, and personal electronics markets; specializing in isolated AC/DC and DC/DC applications. His design and research interests include high-efficiency and high-power-density power conversion, soft-switching converters, and GaN implementation in AC/DC converters. Max obtained a master’s degree in electrical engineering from Zhejiang University in 2006. He has worked at Delta, Power Integrations, Infineon and Texas Instruments.

Related Content

• Ungluing a GaN charger
• USB Type-C PD 3.0 Specification, Charging and Design
• Power Tips #138: 3 ways to close the control loop for totem-pole bridgeless PFC
• Power Tips #127: Using advanced control methods to increase the power density of GaN-based PFC
• Power Tips #115: How GaN switch integration enables low THD and high efficiency in PFC
• Power Tips #131: Planar transformer size and efficiency optimization algorithm for a 1 kW high-density LLC power module

The post Power Tips #139: How to simplify AC/DC flyback design with a self-biased converter appeared first on EDN.